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\fBSUPPLEMENTS\ TO\ THE\ SERIES\ O\ RECOMMENDATIONS\fR \v'2P'
.EF '%	 \ \ \ ^''
.OF ''' \ \ \ ^	%'
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(Section 3 of the Supplements to the Series M, N and O Recommendations)
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\fBMONTAGE:\fR \ PAGE 206 = PAGE BLANCHE
.EF '%	Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.1''
.OF '''Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.1	%'
.sp 1P
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\fB3\ Measuring equipment specifications\fR 
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\fBSupplement\ No.\ 3.1\fR 
.RT
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\fBMEASURING\ INSTRUMENT\ REQUIREMENTS\ \(em\fR 
.EF '%	Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.1''
.OF '''Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.1	%'
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\fBSINUSOIDAL\ SIGNAL\ GENERATORS\ AND\ LEVEL\(hyMEASURING\ INSTRUMENTS\fR  |
.FS
For the convenience of the reader of this Book, this Supplement is republished 
from Volume\ IV.2 of the CCITT \fIGreen\ book\fR , ITU, Geneva, 1973. 
.FE
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\fI(Geneva, 1972; amended at Melbourne, 1988)\fR 
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A.
	\fIDirect\(hyreading, general\(hypurpose, continuously variable\fR 
\fIsinusoidal generator (not sweep frequency)\fR 
.sp 1P
.RT
.PP
Table 1 is a list of the essential performance requirements of a
range of direct reading, general purpose continuously variable sinusoidal
generators.
.PP
If discrete frequencies are required, suitable nominal values for
international purposes are given in Recommendation\ M.580 for telephone\(hytype
circuits and Recommendation\ N.21 for sound\(hyprogramme circuits.
.RT
.sp 2P
.LP
B.
	\fIDirect\(hyreading, general\(hypurpose wideband and selective\fR 
\fIlevel\(hymeasuring instruments (not sweep display or fixed frequency)\fR 
.sp 1P
.RT
.PP
Table 2 is a list of the essential performance requirements of a
range of direct\(hyreading, general\(hypurpose wideband and selective level\(hymeasuring 
instruments. 
.PP
\fINote\fR \ \(em\ The specifications given in\ \(sc\ 8 of Recommendation\ 
O.22 are 
recommended to be used for signal generators and level\(hymeasuring instruments 
to be used on telephone\(hytype circuits. 
.RT
.LP
.sp 22
.bp
.ce
\fBH.T. [T1.3.1]\fR 
.ps 9
.vs 11
.nr VS 11
.nr PS 9
.TS
center box;
cw(342p) .
TABLE\ 1
.T&
cw(342p) .
 {
\fBEssential performance requirements for sinusoidal signal
generators\fR
(not sweep generators)
 }
.TE
.TS
lw(126p) | cw(54p) | cw(54p) | cw(54p) | cw(54p) .
	Telephone\(hytype circuits	Sound\(hyprogramme circuits	 {
Groups, supergroups,
and 12\(hy, 60\(hy, 120\(hy, and
300\(hychannel
systems
 }	 {
Mastergroups, super\(hy
mastergroups and 900\(hy to
2700\(hychannel
systems
 }
_
.T&
cw(126p) | cw(54p) | cw(54p) | cw(54p) | cw(54p) .
1	2	3	4	5
_
.T&
lw(126p) | cw(54p) | cw(54p) | cw(54p) | cw(54p) .
\fIFrequency\fR				
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
a) range	200 Hz to 4 kHz	30 Hz to 20 kHz	4 to 1400 kHz	60 Hz to 17 MHz
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
b)
accuracy of initial setting, without frequency counter,
at
20\ \(deC and nominal power supplies
 }	\(+- | % | (+- |  Hz	\(+- | % | (+- |  Hz	 {
below 120 kHz: \(+- | .2 |  | (+- | 00 Hz
120 kHz land above:
\(+- | .2% | (+- |  kHz
 }	\(+- | .002% | (+- | 00 Hz
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
c) stability				
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
\(em
per hour at 20 | (deC and with nominal power supplies
 }	\(+- | %	\(+- | %	\(+- | .01% | (+- | 50 Hz	\(+- | .005% | (+- | 50 Hz
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
\(em
per 10 | (deC over a specified range of temperature and with
nominal power supplies (Note)
 }	\(+- | .1%	\(+- | .1%	\(+- | .1% | (+- | 50 Hz	\(+- | .002% | (+- | 0 Hz
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
\(em
per 10% change in power suppy at 20 | (deC
 }	\(+- | .5%	\(+- | .5%	\(+- | .05% | (+- | 50 Hz	\(+- | .001% | (+- | 0 Hz
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
\fIOuput level\fR				
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
a) range	 {
+10 to \(em40 dBm
(+12 to \(em45 dNm)
 }	 {
+20 to \(em40 dBm
(+23 to \(em45 dNm)
 }	 {
+10 to \(em60 dBm
(+12 to \(em70 dNm)
 }	 {
+10 to \(em60 dBm
(+12 to \(em70 dNm)
 }
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
b)
accuracy at 0 dBm (0 dNm) and at the reference frequency
at 20 | (deC and with nominal power supplies
 }	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .2 dB (\(+- | .2 dNp)	\(+- | .2 dB (\(+- | .2 dNp)
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
c)
accuracy at any level or frequency within the range
 }	\(+- | .5 dB (\(+- | .6 dNp)	\(+- | .5 dB (\(+- | .6 dNp)	\(+- | .5 dB (\(+- | .6 dNp)	\(+- | .5 dB (\(+- | .6 dNp)
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
d) stability				
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
\(em
per hour at 20 | (deC and with nominal power supplies
 }	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
\(em
per 10 | (deC over a specified range of temperature
and with nominal power supplies (Note)
 }	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
\(em
per 10% change in power supply at 20 | (deC
 }	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
\fIPurity of output\fR				
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
Ratio of total output power to power of unwanted signals (noise, harmonic and non\(hyharmonic frequencies)
 }	at least 40 dB (46 dNp)	at least 50 dB (57 dNp)	at least 46 dB (53 dNp)	at least 46 dB (53 dNp)
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
\fIOutput impedance\fR				
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
a)
nominal value (other values may be specified if
required)
 }	600 ohms balanced	 {
600 ohms balanced or not greater than 6 ohms balanced for constant
voltage techniques
 }	 {
75 ohms unbalanced or 150 ohms balanced or 600 ohms balanced
 }	50 or 75 ohms unbalanced
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
b)
return loss against the nominal value
 }	at least 30 dB (35 dNp)	at least 30 dB (35 dNp)	at least 30 dB (35 dNp)	at least 30 dB (35 dNp)
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
c)
balance about earth (where applicable)
 }	at least 40 dB (46 dNp)	at least 60 dB (70 dNp)	at least 40 dB (46 dNp)	
_
.TE
.nr PS 9
.RT
.ad r
\fBTableau 1 [T1.3.1], p.1 \*`a l'italienne\fR 
.sp 1P
.RT
.ad b
.RT
.LP
.bp
.ce
\fBH.T. [1T2.3.1]\fR 
.ps 9
.vs 11
.nr VS 11
.nr PS 9
.TS
center box;
cw(342p) .
TABLE\ 2
.T&
cw(342p) .
 {
\fBEssential performance requirements of wideband and selective
level\(hymeasuring instruments\fR
(not sweep\(hydisplay of fixed frequency)
 }
.TE
.TS
lw(126p) | cw(54p) | cw(54p) | cw(54p) | cw(54p) .
	Telephone\(hytype circuits	Sound\(hyprogramme circuits	 {
Groups, supergroups,
and 12\(hy, 60\(hy,120\(hy and
300\(hychannel
systems
 }	 {
Mastergroups, super\(hy
mastergroups and 900\(hy to
2700\(hychannel
systems
 }
_
.T&
cw(126p) | cw(54p) | cw(54p) | cw(54p) | cw(54p) .
1	2	3	4	5
_
.T&
lw(126p) | cw(54p) | cw(54p) | cw(54p) | cw(54p) .
\fIFrequency\fR				
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
a) range	200 Hz to 4 kHz	30 Hz to 20 kHz	4 kHz to 1400 kHz	60 kHz to 17 MHz
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
b)
nominal bandwith for selective measurements
(Note 1)
 }	40 Hz	40 Hz	600 Hz and 4 kHz	600 Hz and 4 kHz
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
\fIRange of input level\fR
 }				
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
a) wideband	 {
+20 to \(em50 dBm
(+23 to \(em58 dNm) down to \(em70 dBm (\(em80 dNm)
with reduced accuracy
 }	 {
+20 to \(em50 dBm
(+23 to \(em58 dNm) down to \(em70 dBm (\(em80 dNm)
with reduced accuracy
 }	 {
+20 to \(em50 dBm
(+23 to \(em58 dNm)
 }	 {
+20 to \(em50 dBm
(+23 to \(em58 dNm)
 }
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
b) selective	 {
+20 to \(em80 dBm
(+23 to \(em92 dNm)
 }	 {
+20 to \(em80 dBm
(+23 to \(em92 dNm)
 }	 {
+20 to \(em90 dBm
(+23 to \(em100 dBm) down to \(em110 dBm
(\(em127 dNm)
with reduced accuracy
 }	 {
+20 to \(em90 dBm
(+23 to \(em100 dBm) down to \(em110 dBm\fR
(\(em127 dNm)
with reduced accuracy
 }
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
\fIMeasuring accuracy\fR				
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
a)
at 0 dBm (0 dNm) and at the reference frequency at 20 | (deC
and with nominal power supplies if internal calibration is provided
 }	 {
\(+- | .2 dB (\(+- | .2 dNp)
\(+- | .1 dB (\(+- | .1 dNp)
 }	 {
\(+- | .2 dB (\(+- | .2 dNp)
\(+- | .1 dB (\(+- | .1 dNp)
 }	 {
\(+- | .2 dB (\(+- | .2 dNp)
\(+- | .1 dB (\(+- | .1 dNp)
 }	 {
\(+- | .2 dB (\(+- | .2 dNp)
\(+- | .1 dB (\(+- | .1 dNp)
 }
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
b)
at any level and frequency within the ranges
(Note\ 2)
 }	\(+- | .5 dB (\(+- | .6 dNp)	\(+- | .5 dB (\(+- | .6 dNp)	\(+- | .5 dB (\(+- | .6 dNp)	\(+- | .5 dB (\(+- | .6 dNp)
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
\fIStability of indicated level\fR
(Note\ 3)
 }				
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
a)
per hour at 20 | (deC and with nominal power supplies
 }	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
b)
per 48 hours at 20 | (deC and with nominal power supplies
 }	\(+- | .3 dB (\(+- | .4 dNp)	\(+- | .3 dB (\(+- | .4 dNp)	\(+- | .3 dB (\(+- | .4 dNp)	\(+- | .3 dB (\(+- | .4 dNp)
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
c)
per 10 | (deC over a specified range of temperature and with nominal
power supplies (Note\ 3)
 }	\(+- | .5 dB (\(+- | .6 dNp)	\(+- | .5 dB (\(+- | .6 dNp)	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
d)
per 10% change in power supply at
20 | (deC
 }	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)	\(+- | .1 dB (\(+- | .1 dNp)
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
\fILevel of unwanted signals\fR
 }				
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
Generated by the instrument itself and appearing at the input
terminals relative to the lowest acceptable input level measured 
at\ the
input terminals
 }	 {
\(em20 dB (\(em23 dNp)
or lower
 }	 {
\(em20 dB (\(em23 dNp)
or lower
 }	 {
\(em20 dB (\(em23 dNp)
or lower
 }	 {
\(em20 dB (\(em23 dNp)
or lower
 }
_
.TE
.nr PS 9
.RT
.ad r
\fBTableau 2 [1T2.3.1], p.\fR 
.sp 1P
.RT
.ad b
.RT
.LP
.bp
.ce
\fBH.T. [2T2.3.1]\fR 
.ps 9
.vs 11
.nr VS 11
.nr PS 9
.TS
center box;
cw(342p) .
TABLE\ 2 \fI(cont.)\fR
.TE
.TS
lw(126p) | cw(54p) | cw(54p) | cw(54p) | cw(54p) .
	Telephone\(hytype circuits	Sound\(hyprogramme circuits	 {
Groups, supergroups,
and 12\(hy, 60\(hy, 120\(hy and
300\(hychannels
systems
 }	 {
Mastergroups, super\(hy
mastergroups and 900\(hy to
2700\(hychannel
systems
 }
_
.T&
cw(126p) | cw(54p) | cw(54p) | cw(54p) | cw(54p) .
1	2	3	4	5
_
.T&
lw(126p) | cw(54p) | cw(54p) | cw(54p) | cw(54p) .
\fIInput impedance\fR				
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
a)
nominal value for terminated level measurements.
Other
nominal values may be specified if required
 }	600 ohms balanced	 {
600 ohms balanced or at least 20 | (mu | 0\u3\d ohms balanced for constant
voltage techniques
 }	 {
75 ohms unbalanced or 150 or 600 ohms either balanced or unbalanced
 }	50 or 75 ohms unbalanced
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
b)
value for through\(hylevel measurements
 }	 {
at least 25 | (mu | 0\u3\d ohms balanced
 }	 {
at least 20 | (mu | 0\u3\d ohms
 }	 {
through\(hylevel measurements not recommended
 }	 {
through\(hylevel measurements not recommended
 }
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
c)
return loss against nominal value (for terminated\(hylevel
measurements)
 }	at least 30 dB (35 dNp)	at least 30 dB (35 dNp)	at least 30 dB (35 dNp)	at least 30 dB (35 dNp)
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
d)
balance about earth where applicable through\(hylevel or
terminated\(hylevel
 }	at least 40 dB (46 dNp)	at least 60 dB (70 dNp)	at least 40 dB (46 dNp)	
.T&
lw(126p) | lw(54p) | lw(54p) | lw(54p) | lw(54p) .
 {
\fIImage frequency rejection\fR
 }	at least 50 dB (58 dNp)	at least 50 dB (58 dNp)	at least 60 dB (70 dNp)	 {
at least 60 dB (70 dNp)
 }
.TE
.LP
\fINote\ 1\fR
\ \(em\ It is necessary to specify in some detail the response
characteristic of the nominal bandwidth for selective measurements.
.LP
\fINote\ 2\fR
\ \(em\ Although the actual return loss of the input impedance is specified to be not greater than 30\ dB (35\ dNp) the instrument should be arranged (when connected to a generator of exactly the appropriate nominal value) to indicate the level that would be developed across an impedance, with a return loss of at least 40\ dB (46\ dNp) against the nominal value.
.LP
\fINote\ 3\fR
\ \(em\ The stability limits include the effects of frequency variation of any built\(hyin local oscillator in selective measuring sets.
.LP
\fINote\ 4\fR
\ \(em\ The range of temperature over which the apparatus must
satisfactorily operate must be specified. This depends very largely on
geographical location.
.nr PS 9
.RT
.ad r
\fBTableau 2 (fin) [2T2.3.1], p.\fR 
.sp 1P
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.RT
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.bp
.sp 2P
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\fBSupplement\ No.\ 3.2\fR 
.RT
.sp 2P
.sp 1P
.ce 1000
\fBNOISE\ MEASURING\ INSTRUMENTS\ FOR\ TELECOMMUNICATION\ CIRCUITS\fR 
.EF '%	Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.2''
.OF '''Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.2	%'
.ce 0
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(For this Supplement, see page 534, Volume IV.2 of the
\fIGreen\ Book\fR )
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\fBSupplement\ No.\ 3.3\fR 
.RT
.sp 2P
.sp 1P
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\fBPRINCIPAL\ CHARACTERISTICS\ OF\ VOLUME\ INDICATORS\fR 
.EF '%	Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.3''
.OF '''Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.3	%'
.ce 0
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(For this Supplement, see page\ 548, Volume IV.2 of
the \fIGreen\ Book\fR ;
.sp 9p
.RT
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additional information on this subject
is given in Recommendation\ O.51)
\v'2P'
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.sp 2P
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\fBSupplement\ No.\ 3.4\fR 
.RT
.sp 2P
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\fBCONSIDERATION OF INTERWORKING BETWEEN DIFFERENT\fR 
.EF '%	Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.4''
.OF '''Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.4	%'
.ce 0
.sp 1P
.ce 1000
\fBDESIGNS OF APPARATUS FOR MEASURING QUANTIZING DISTORTION\fR 
.ce 0
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(For this Supplement, see page 85, Volume IV.2 of
the \fIOrange\ Book\fR )
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\fBSupplement\ No.\ 3.5\fR 
.RT
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(Cancelled. Replaced by Recommendation O.6)
\v'2P'
.EF '%	Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.5''
.OF '''Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.5	%'
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\fBSupplement\ No.\ 3.6\fR 
.RT
.sp 2P
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\fBCROSSTALK\ TEST\ DEVICE\ FOR\ CARRIER\(hyTRANSMISSION\fR 
.EF '%	Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.6''
.OF '''Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.6	%'
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\fBON\ COAXIAL\ SYSTEMS\fR 
.ce 0
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\fI(Melbourne, 1988)\fR 
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(Information from the USSR Telecommunication Administration)
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.LP
\fB1\fR 	\fBIntroduction\fR 
.sp 1P
.RT
.PP
This Supplement contains the description of a method and the basic technical 
parameters of a device for 
crosstalk ratio measurement
. It is designed for remote localization of repeaters having a low near\(hyend
intelligible crosstalk ratio in carrier\(hytransmission coaxial systems.
.bp
.RT
.sp 2P
.LP
\fB2\fR 	\fBOperation\fR 
.sp 1P
.RT
.PP
The device measures 
propagation delay time of near\(hyend
crosstalk signals
from different repeaters. Measurement of the test signal delay time in 
order to determine the distance from a repeater and the amplitude of the 
received signal make it possible to determine the repeater number and 
the near\(hyend crosstalk ratio of this repeater.
.PP
The test signal is extracted from the noise and signals, coming from other 
repeaters, by means of time filtering (correlation processing). It is 
preferred that a special signal having a sufficiently narrow correlation
function be used as a test signal. A sinusoidal test signal phase\(hymodulated 
by a pseudorandom sequence (PRS) of pulses (phase\(hymodulated signal) 
is used in the device. 
.PP
A simplified block diagram and a frequency diagram of this device are given 
in Figure\ 1 and Figure\ 2. 
.PP
Phase modulation of a sinusoidal signal
\fIf\fR\d1\ufrom an oscillator G1 by a signal from PRS oscillator G2 is
carried out in a modulator M1, the formed signal spectrum having no spectral
component \fIf\fR\d1\u(suppressed by more than 54\ dB). The modulating and test
signals are shown in Figure\ 3, and the modulating signal spectrum is shown 
in Figure\ 4. A phase\(hymodulated test signal in the band from \fIf\fR\d2\u\fI\fI\d\fIm\fR\uto 
\fIf\fR\d\fIk\fR\\d\fIm\fR\uis formed in a modulator M3. A signal from 
a quartz controlled 
oscillator at one of the frequencies in the band from \fIf\fR\d2\uto
\fIf\fR\d\fIk\fR\u, which
are chosen in the spectrum of transmission systems under test, is used as a
carrier. A test signal at \fIf\fR\d\fIk\fR\\d\fIm\fR\u\ \(+-\ \fIf\fR\d1\u\fI\fI\d\fIm\fR\uas 
well as at 
\fIf\fR\d1\u\fI\fI\d\fIm\fR\ucontain no central spectral component. The 
signal \fIf\fR\d\fIk\fR\\d\fIm\fR\uis applied to the input of an interfering 
link. 
.PP
A crosstalk signal from the output of the return path (path subjected to 
interference) is applied to the input of the device. The signal is 
reconverted in modulator M4. The signal \fIf\fR\d1\u\fI\fI\d\fIm\fR\uis 
then applied to an 
input of phase detector M2. The PRS signal from G2 shifted by the time 
interval of \(*D"\fIt\fR with respect to the modulating signal in a time\(hydelay 
circuit D1 is 
applied to the other input of the phase detector M2. If the present time
interval coincides with the time delay of the crosstalk signal in a line 
being tested with respect to the test signal at the device output, a single\(hyfrequency 
sinusoidal signal \fIf\fR\d1\uwill be obtained at the output of M2, the 
signal 
level then being measured by a selective level meter (SLM). When the present 
value of \(*D"\fIt\fR does not coincide with the time delay of the crosstalk 
signal 
coming from the line, a signal having no frequency \fIf\fR\d1\uin its spectrum 
will be present at the output and input of the phase detector M2. By varying 
the 
value of the present time delay in D\d1\u, tuning to a crosstalk signal from
different repeaters on the section under test, a remote measurement of the
crosstalk value of all repeaters is carried out.
.PP
It is preferred that the choice of parameters of the test signal be
determined by the correlation function \fIR\fR (\fIt\fR ) of the chosen 
signal (see 
Figure\ 5). For this purpose, \fIR\fR (\fIt\fR ) is estimated at two levels:
\fIR\fR (\fIt\fR )\ \(=\ 0.1 corresponding to the zone of low correlation and
\fIR\fR (\fIt\fR )\ =\ 0.607 limiting the high correlation zone.
.PP
Resolution between two adjacent signals is practical if the time
shifts between them is outside the zone of high correlation. Therefore, the
choice of the duration of an elementary PRS pulse is made depending on the
minimum crosstalk time shift \(*D"\fIt\fR\d\fIm\fR\\d\fIi\fR\\d\fIn\fR\uof 
crosstalk from the adjacent 
repeaters, namely:
\v'6p'
.RT
.ad r
.ad b
.RT
.LP
where
.LP
	\fIl\fR\d\fIR\fR\\d\fIS\fR\u	is the minimum distance between the adjacent
repeaters;
.LP
	\fIV\fR 	is the electric wave propagation rate in the cable.
.PP
The pulse duration \(*t in the device depends on the scale oscillator frequency 
and may be adjusted for various cable types having different 
propagation rates. Adjustment is carried out by changing the scale oscillator 
frequency. 
.PP
The repetition period of a pseudorandom sequence should ensure
unambiguity of measurements, i.e. the time between two adjacent autocorrelation 
function maximums should be greater than the signal propagation time along 
the section \fIl\fR\d\fIS\fR\\d\fIT\fR\uunder test in both directions of 
transmission: 
\v'6p'
.RT
.ad r
.ad b
.RT
.LP
.bp
.PP
The minimum step of the time\(hydelay circuit D1 is determined by
taking into account the admissible error of tuning to the maximum of the
autocorrelation function and may be equal to 0.1\ \(*t (error not more 
than 5%). 
The maximum value of the time delay in D1 is determined by the length of the
line section \fIl\fR\d\fIS\fR\\d\fIT\fR\uunder test, i.e. by the time of 
signal propagation 
along the line in both directions of transmission:
\v'6p'
.ad r
.ad b
.RT
.PP
To measure the crosstalk signal levels corresponding not only to low but 
also normal crosstalk attenuation of repeaters, the passband of the SLM 
must be sufficiently narrow (0.1 to 0.3\ Hz) so that a test signal may 
be 
extracted from the noise. Such a passband may be realized by means of a
synchronous phase filter.
.sp 2P
.LP
\fB3\fR 	\fBBasic technical parameters of a device designed for transmission
systems at frequencies less than 18\ MHz\fR 
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3.1
	\fIBasic characteristics\fR 
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3.1.1
	Maximum length of a section under test
\ 400\ km
.LP
3.1.2
	Minimum distance between repeaters under test
\ 1.0\ km
.LP
3.1.3
	Minimum step of setting distance to the repeater under
test
\ 0.1\ km
.ad r
3.1.4
	Nominal carrier frequencies of a test signal
\ 0.37; 1.1;
\ 4.4; 7.9;
\ 17.25\ MHz
.ad b
.RT
.LP
	3.1.5
	Minimum measurement level
\ \(em120\ dB
.LP
3.1.6
	Time for localization of a faulty repeater
.LP
(with a maximum of 70 repeaters on a section under test)
\ 20\ min
.sp 1P
.LP
3.2
	\fISeveral technical characteristics\fR 
.sp 9p
.RT
.LP
3.2.1
	Number of elementary pulses in a pseudorandom sequence (PRS)
.LP
for the test signal phase modulation
\ 2\u9\d\ \(em\ 1\ =\ 511
.LP
3.2.2
	PRS repetition rate
\ 4.2\ ms
.LP
3.2.3
	Test signal level range
\ \(em59\ dB to 0\ dB
.LP
3.2.4
	Scale oscillator frequency
\ 2.4 to 2.5 MHz
.LP
3.2.5
	Level measuring range
\ \(em120 to \(em50\ dB
.LP
3.2.6
	Receiver bandwidth (at a 3\ dB level)
\ 0.3; 3\ Hz
.ad r
3.2.7
	Steps of time delay
\ 83.3\ \(*ms (10\ km)
\ 8.3\ \(*ms (1\ km)
\ 0.8\ \(*ms (0.1\ km)
.ad b
.RT
.LP
	3.2.8
	Reduction in the receiver indicator reading with respect to a value
corresponding to the maximum when the PRS is shifted by 24.9\ \(*ms
(3\ km)
\ more than 40\ dB
.LP
3.2.9
	Measuring error in the \*Q\(em100\ dB\*U range for the 0\ dB
reading
\ less than \(+- | \ dB
.LP
.rs
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BLANC
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.bp
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.rs
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\fBFigure 1, (N), p. 4\fR 
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\fB
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\fBFigure 2, (N), p. 5\fR 
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\fB
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\fBFigure 3, (N), p. 6\fR 
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\fB
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\fBFigure 4, (N), p. 7\fR 
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\fB
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.rs
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\fBFigure 5, (N), p. 8\fR 
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\fB
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\fBSupplement\ No.\ 3.7\fR 
.RT
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.ce 1000
\fBA\ MEASURING\ SIGNAL\ (MULTITONE\ TEST\ SIGNAL)\ FOR\ FAST\ MEASUREMENT\fR 
.EF '%	Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.7''
.OF '''Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.7	%'
.ce 0
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\fBOF\ AMPLITUDE\ AND\ PHASE\ FOR\ TELEPHONE\ TYPE\ CIRCUITS\fR 
.ce 0
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\fI(Melbourne, 1988)\fR 
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(Information submitted by the Federal Republic of Germany,
France and USSR)
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.PP
In the following a brief description of a test signal is given,
stating its particular advantages for measurement of amplitude and phase
simultaneously.
.sp 1P
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\fB1\fR 	\fBThe multitone test signal\fR 
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1.1
	\fIGeneral description\fR 
.sp 9p
.RT
.PP
The multitone test signal (MTTS) consists of a spectrum of N
discrete signals separated by frequency spacing of 100\ Hz in the low frequency 
range. 
.PP
The spectral lines are all of equal amplitude; their phase
relationship to each other is chosen on the basis of mathematical
considerations so that the energy of the test signal is distributed
approximately evenly across the entire period of the test signal.
.bp
.PP
The transmission characteristics, i.e. the amplitude and phase
distortion of a telephone line, produce changes in the test signal. On the
receive side, these changes are measured and evaluated, e.g. by means of a
Fourier analysis. The results may be displayed on a screen in the form of an
amplitude and/or phase graph and also, for example, the group delay may be
derived from this.
.RT
.sp 1P
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1.2
	\fIMeasuring principle\fR 
.sp 9p
.RT
.PP
The transmit signal consisting of N cosine waveforms is generated in digital 
circuits: a sufficient number of instantaneous values of the MTTS is read 
out of a ROM with a clock frequency. After passing through a D/A converter 
and a filter which suppresses the clock frequency, the composite signal 
is 
available:
\v'6p'
.RT
.ad r
.ad b
.RT
.LP
where
.LP
	\fIA\fR 	amplitude of a single waveform
.LP
	\fIf\fR 	is 100 Hz (see Note\ 2)
.LP
	\(*f
	phase of the single waveforms
.LP
	\fIn\fR 	serial number of the single waveforms
.LP
	\fIt\fR 	time
.LP
	\fIN\fR 	total number of waveforms.
.PP
At \fIf\fR = 100 Hz, the duration of one period of the MTTS is
10\ ms.
.PP
The MTTS is passed to the object to be tested which changes the
properties of the MTTS, i.e. the amplitudes and phases of the single
waveforms.
.PP
In the receiving section, the changed signal is passed to an
evaluation circuit, where the signal is sampled with the clock frequency. 
The sampled analogue values are digitized and stored in a memory. The stored 
values of the time function are then transferred by means of the Discrete 
Fourier 
Transform into the frequency domain. All necessary calculations are performed 
in a microcomputer. 
.PP
At measurements where the objects to be tested include carrier
frequency systems, frequency shift of the measuring signal can appear. 
In such cases it is recommended to use window functions in the signal processing 
section of the receiver.
.PP
The characteristics of the object to be tested are derived from the
deviation of the received values against the transmitted values.
.RT
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1.3
	\fIData of the multitone test signal\fR 
.sp 9p
.RT
.PP
\fITransmitter\fR 
.PP
Transmit frequencies
.RT
.LP
	\(em
	35 signals (cosine) simultaneously;
.LP
	\(em
	\fIn\fR \(mu 100 Hz; \fIn\fR = 2 to 36 in steps of 100 Hz from 200 to
3600\ Hz, or see Notes\ 1 and\ 2;
.LP
	\(em
	Accuracy: 1\ \(mu\ 10\uD\dlF261\u4\d
.PP
Transmit level (multitone test signal) +10 to \(em40\ dBm.
.PP
This level corresponds to the level of a single sinusoidal signal
which has the same peak value as the test signal.
.RT
.LP
	\(em
	Accuracy at 1000\ Hz
	0.2\ dB
.LP
	\(em
	Frequency response
	0.1\ dB
.LP
	\(em
	Harmonic distortion
	40\ dB
.LP
	\(em
	Spurious distortion at +10\ dBm
	50\ dB
.LP
	\(em
	Phase constellation
.bp
.ce
\fBH.T. [T1.3.7]\fR 
.ps 9
.vs 11
.nr VS 11
.nr PS 9
.TS
center box;
cw(35p) | cw(35p) | cw(29p) | cw(35p) | cw(29p) | cw(35p) | cw(30p) .
0	2\(*p/7	4\(*p/7	6\(*p/7	8\(*p/7	10\(*p/7	12\(*p/7
_
.T&
lw(10p) | lw(25p) | lw(35p) | lw(29p) | lw(35p) | lw(29p) | lw(35p) | lw(30p) .
n:	 {
2, 3, 4, 5, 6, 8, 15, 22, 29, 36
 }	9, 12, 20, 24, 35	 {
10, 16, 18, 26, 28, 34
37 (Note 1)
 }	11, 13, 31, 33	21, 23, 27, 32 1 (Note 1)	14, 19, 25, 30	 {
7, 17
38 (Note 1)
 }
.TE
.LP
\fINote\ 1\fR
\ \(em\ Serial numbers of 1, 37 and 38 are optional values.
.LP
\fINote\ 2\fR
\ \(em\ The French Administration uses frequency steps of
101.56\ Hz according to [26 | (mu | \fIn\fR
\(em1)] | (mu | fIf\fR
, where \fIf\fR
\ =\ 8000/2048.
This is in accordance with the principle of frequency offset contained in
Recommendation\ O.6 concerning PCM equipment.
.nr PS 9
.RT
.ad r
\fBTableau [T1.3.7], p.9\fR 
.sp 1P
.RT
.ad b
.RT
.PP
\fIReceiver\fR 
.PP
The receiver takes into account the level and the phase constellation of 
the transmitted signal. 
.RT
.sp 2P
.LP
\fB2\fR 	\fBAdvantages of the multitone test signal\fR 
.sp 1P
.RT
.PP
With the technical means available today the multitone test signal can 
be generated at low cost with excellent stability of frequency, amplitude 
and phase. The quantity of\ 35 discrete signals and thus test points in 
the 
frequency range\ 200 to 3600\ Hz is quite adequate for the testing requirements 
occurring in practice. Optionally, the frequency band can be widened according 
to Note\ 1. 
.PP
When the received signal is evaluated, e.g. with the aid of a Fourier analysis 
to determine amplitude/frequency response and/or phase or group delay, 
a test cycle time, allowing for processing time and screen display time, 
of 
only less than one second is needed. This short test cycle is of great
advantage mainly when equalization work has to be done.
.PP
Because the MTTS is normally a continuous signal there are no settling 
time problems which occur using a sweep mode signal. 
.PP
The MTTS is an ideal band\(hylimited \*Qnoise signal\*U for determining 
the rms bandwidth of filters, for example for the filter (psophometric 
weighting) in Recommendation\ O.41 or for calibrating PCM instruments measuring 
quantizing distortion. 
.PP
Considering the ripple at the frequency response curve one can
recognize very clearly that there are frequency components caused by any
non\(hylinearity of an item under test.
.PP
Using the Fourier analysis to evaluate the received MTTS one can
recognize both the amplitude and frequency of unwanted signals; that means, 
the procedure works like a swept selective receiver. 
.PP
The period of this MTTS is 10\ ms (which corresponds to one period of a 
100\(hyHz fundamental). Since for Fourier analysis it is sufficient to 
sample just one period of the test signal, i.e. 10\ ms, at the receiving 
side, and 10\ ms 
plus at the sending side, measurements could be performed during
correspondingly short gaps in the speech or data transmission signal. These
gaps occur in any case in these signals, or they may be created by technical
means.
.PP
The use of the MTTS in combination with the Fourier analysis makes it possible 
to provide measurements of parameters which normally require filters; e.g. 
weighted noise, quantizing distortion, selective crosstalk, etc. In these 
cases filtering is provided by appropriate calculations in the microcomputer 
carrier out for the frequency domain of the input signal.
.PP
For measurements including PCM sections it is not necessary to shift the 
frequencies in order to avoid submultiples of\ 8\ kHz, in this case a MTTS 
without frequency shift leads into a frequency response with a ripple of 
up to \(+-\ 0.1\ dB. With the help of an averaging procedure (e.g.\ 4 or\ 
16 measuring 
cycles) the ripple can be reduced to a negligible value.
.bp
.PP
A further possibility to reduce the ripple is to use shifted
frequencies of \fIn\fR  | (mu | 01 | (mu | 6\ Hz, according to Note\ 2.
.PP
In this case the ripple is less than \(+- | .05\ dB after one measuring
cycle; even this relatively small error can be reduced by an averaging
procedure.
.RT
.sp 2P
.LP
\fB3\fR 	\fBPractical experience\fR 
.sp 1P
.RT
.PP
Since 1981, instruments using multitone test signals have been used by 
various Administrations all over the world. 
.PP
Measurement results are obtained quickly and unambigously and are
compatible with those obtained with conventional methods.
.PP
The USSR Telecommunication Administration is investigating
theoretically and practically the MTTS in order to determine the best use 
for further applications. 
.RT
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\fBSupplement\ No.\ 3.8\fR 
.RT
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.sp 1P
.ce 1000
\fBGUIDELINES\ CONCERNING\ THE\ MEASUREMENT\ OF\ JITTER\fR 
.EF '%	Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.8''
.OF '''Fascicle\ IV.4\ \(em\ Suppl.\ No.\ 3.8	%'
.ce 0
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\fI(Melbourne, 1988)\fR 
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.ce 0
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.ce 1000
(Information assembled by SG IV and SG XVIII)
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.ce 0
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\fB1\fR 	\fBDefinitions and causes of jitter\fR 
.sp 1P
.RT
.PP
CCITT Recommendation\ G.701 [1] defines jitter as \*Qshort\(hyterm
non\(hycumulative variations of the significant instants of a digital signal 
from their ideal position in time\*U. This means that jitter is an (unwanted) 
phase 
modulation of the digital signal. The frequency of the phase variations is
calle 
jitter\(hyfrequency
. A second parameter which is closely related to jitter is called wander. 
It is defined as \*Qlong\(hyterm non\(hycumulative variations of the significant 
instants of a digital signal from their ideal position in 
time\*U. Up to now there is no clear definition of the boundary between jitter
and wander. Components of phase variation having frequencies below the range
of\ 1 to\ 10\ Hz are normally called wander.
.PP
Jitter may deteriorate the transmission performance of a digital
circuit. As a result of signal displacement from its ideal position in time,
errors may be introduced into the digital bit stream at points of signal
regeneration. Slips may be introduced into digital signals resulting from
either data overflow or depletion in digital equipment incorporating buffer
stores and phase comparators. In addition, phase modulation of the
reconstructed samples in digital\(hyto\(hyanalogue conversion devices may 
result in degradation of the decoded analogue signals. This is more likely 
to be a 
problem when transmitting encoded wide\(hyband signals.
.PP
A distinction must be made between the systematic and random jitter.
Systematic jitter
results from misaligned timing recovery circuits in signal regenerating 
devices or from inter\(hysymbol interference and 
amplitude\(hyto\(hyphase conversion caused by imperfect cable equalization.
Systematic jitter is pattern\(hydependent.
.PP
Random jitter
originates from internal or external
interferring signals such as repeater noise, crosstalk or reflections. 
Random jitter is independent of the transmitted pattern. 
.PP
Low\(hyfrequency jitter
produced in pulse justification
demultiplexers arises from pulse justification synchronization; the mechanism 
by which the plesiochronous lower\(hyrate signals are synchronized to a 
locally 
generated clock source. This jitter, which appears at the demultiplexer
lower\(hyrate output, is denoted \*Q
justification jitter
\*U or \*Q
waiting
time jitter
\*U.
.PP
As systematic jitter is correlated with the transmitted pulse pattern at 
different regenerators, it accumulates coherently. Random jitter is 
uncorrelated at different regenerators and accumulates incoherently. In most
existing lower\(hyrate digital systems, systematic jitter is dominant. In some
contemporary higher\(hyrate systems the random component may become significant 
or even predominant. 
.bp
.PP
Unlike some other impairments, disturbing jitter can be reduced by
regenerators or by the use of \*Q
de\(hyjitterizers
\*U which contain a signal buffer with a narrow\(hyband phase\(hysmoothing 
circuit. Regenerators can only reduce jitter frequency components above 
the cut\(hyoff frequency of the clock recovery circuits. At lower jitter 
frequencies, the output signal or a regenerator 
follows the input jitter. In this case jitter is \*Qtransferred\*U which
means that a regenerator behaves like a low\(hypass filter. This characteristic
behaviour leads to the typical jitter tolerance templates as shown in
Figure\ 1.
.RT
.LP
.rs
.sp 33P
.ad r
\fBFigure 1, (N), p.\fR 
.sp 1P
.RT
.ad b
.RT
.PP
It can be seen from the considerations above that jitter can
severely deteriorate the performance of digital transmission systems. On the
other hand, jitter cannot be avoided completely. To evaluate whether jitter 
is kept within the allowed limits is the task of jitter measurements. 
.sp 2P
.LP
\fB2\fR 	\fBTest environment\fR 
.sp 1P
.RT
.PP
In order to facilitate repeatable and accurate measurements, and to allow 
comparisons between measurements made at different times, it is necessary 
to minimize variations in the test environment. Several test environment 
parameters which may vary widely within their allowed ranges and may
significantly affect jitter measurement results (depending upon the type of
equipment involed) include the data pattern, data rate, pulse shape, and 
cable characteristics. The characteristics of these parameters should be 
controlled as appropriate. Additionally, there are secondary test environment 
parameters which may also affect jitter performance, that should be maintained 
at nominal levels to facilitate repeatable measurements. 
.bp
.PP
In order to verify worst\(hycase equipment performance, it may be
necessary to stress the equipment under test with multiple changes in the 
test environment. However, this type of test does not necessarily provide 
meaningful jitter performance data due to lack of control of the particular 
parameter(s) which may be causing errors, as well as their effect on other 
non\(hyjitter 
related equipment failure mechanisms. Therefore, multiple changes in test
environment should not be used to characterize the jitter performance of the
equipment under test.
.RT
.sp 1P
.LP
2.1
	\fIControlled data patterns\fR 
.sp 9p
.RT
.PP
Some measurement procedures require the application of controlled data 
patterns. When the controlled data pattern is intended to approximate live 
traffic encountered in the network, a pseudo\(hyrandom bit sequence (PRBS) 
is 
recommended. Four pseudo\(hyrandom patterns are specified in Recommendations\ 
O.151 and\ O.152 namely 2\u1\d\u1\d\ \(em\ 1, 2\u1\d\u5\d\ \(em\ 1, 2\u2\d\u0\d\ 
\(em\ 1 and 2\u2\d\u3\d\ \(em\ 1 length sequences. To ensure that a particular 
PRBS will generate adequate 
jitter spectral line density within the jitter half\(hypower bandwidth 
of typical clock recovery circuits at the applicable hierarchical level, 
the PRBS word 
length should be much greater than the data rate divided by the jitter
half\(hypower bandwidth. The CCITT recommends that the PRBS word length be at
least 100 times greater than the data rate divided by the jitter half\(hypower
.PP
bandwidth [2] (see Note). The pseudo\(hyrandom bit sequence of 2\u1\d\u5\d\ 
\(em\ 1 bit length specified in Recommendation\ O.151 for bit error measurements 
may 
generate an inadequate spectral line density for jitter measurements a 
speeds above the primary rate. Moreover, this pattern has poor binary run 
properties. Therefore, for bit rates at and above the primary rate, the 
pattern length 
should be no less than 2\u2\d\u0\d\ \(em\ 1, and have a well balanced binary 
run 
characteristic\ [3].
.PP
\fINote\fR \ \(em\ Further study of jitter spectral line density sufficiency 
is desirable. 
.RT
.sp 1P
.LP
2.2
	\fIBit rate\fR 
.sp 9p
.RT
.PP
The bit rate must be maintained within the specifications for
digital interfaces as specified in Recommendation\ G.703 [4]. For convenience, 
the bit rates are repeated below: 
.RT
.LP
	\(em
	basic rate:
	\ \ \  |  64\ kbit/s
.LP
	\(em
	primary rate:
	\ \ 1 | 44\ kbit/s\ \(+-\ 50\ ppm
.LP
	\ \ 2 | 48\ kbit/s\ \(+-\ 50\ ppm
.LP
	\(em
	secondary rate: 
	\ \ 6 | 12\ kbit/s\ \(+-\ 30\ ppm
.LP
	\ \ 8 | 48\ kbit/s\ \(+-\ 30\ ppm
.LP
	\(em
	tertiary rate:
	\ 32 | 64\ kbit/s\ \(+-\ 10\ ppm
.LP
	\ 34 | 68\ kbit/s\ \(+-\ 20\ ppm
.LP
	\ 44 | 36\ kibt/s\ \(+-\ 20\ ppm
.LP
	\(em
	quaternary rate:
	139 | 64\ kibt/s\ \(+-\ 15\ ppm
.sp 1P
.LP
2.3
	\fIPulse shape and cable characteristics\fR 
.sp 9p
.RT
.PP
Pulse shape affects jitter performance by impacting the accuracy of the 
decision making process in a block recovery circuit. Pulse shape is 
typically specified by a pulse template at an output interface or at a
cross\(hyconnect [5] and may vary at the equipment input due to cable effects,
resulting from operating within the specified range of cable lengths and
specified cable type(s). It is recommended that the pulse shape to be used 
in jitter tests be centered within the pulse template specified, rather 
than being at the extreme allowable values (see Note). 
.PP
\fINote\fR \ \(em\ A pulse template appropriate for jitter testing needs 
further study. 
.RT
.sp 1P
.LP
2.4
	\fISecondary test environment parameters\fR 
.sp 9p
.RT
.PP
Other test environment parameters which may affect jitter
performance include temperature, cross\(hytalk, and noise. Temperature affects
jitter performance by altering the resonant fequency of clock recovery
circuits, oscillators, and phase
.bp
.PP
smoothing circuits, as well as
changing
the filtering properties of analog circuitry. Cross\(hytalk may affect jitter
performance when signals in a cable, backplane, or circuit board affect one
another to a noticeable degree. Noise affects the decision making process 
in a clock recovery circuit by decreasing the decision eye margin. 
.PP
In order to obtain accurate and repeatable jitter measurements and
ensure that the effects of jitter applied to the quipment dominate measurement 
results, it is recommended that these secondary parameters be maintained 
at 
their nominal levels.
.RT
.sp 2P
.LP
\fB3\fR 	\fBGlossary or test configuration functional block components\fR 
.sp 1P
.RT
.PP
This glossary defines the functional block components employed in the test 
configurations described in the following sections. Note that these 
functional blocks may be incorporated in various combinations within different 
test equipment. 
.RT
.LP
	\(em
	\fIAttenuator:\fR  | A device which reduces the amplitude of a
digital signal in order to decrease the signal\(hyto\(hynoise ratio.
.LP
	\(em
	 \fIDigital signal generator:\fR  | A signal source which provides a digital 
network hierarchical signal at the appropriate bit rate with proper 
output impedance, pulse shape, line coding, and frame format. This functional 
block component is capable of providing several data patterns, must have 
a 
clock and data output, and may accept an external clock input.
.LP
	\(em
	 \fIDigital signal receiver:\fR  | An instrument which terminates a digital 
network hierarchical signal and monitors for bit errors, errored 
seconds, or bit error ratio (BER).
.LP
	\(em
	\fIEquipment under test (EUT):\fR  | A circuit or system that
is being tested with a controlled data pattern.
.LP
	\(em
	\fIFrequency synthesizer:\fR  | An extremely stable frequency
source of high accuracy. Some frequency synthesizers are capable of adding
phase or frequency modulation (PM or FM) to the primary output while providing 
an unmodulated secondary output. 
.LP
	\(em
	\fIJitter generator:\fR  | An instrument which produces a
hierarchical rate clock modulated by sinusoidal jitter of adjustable frequency 
and amplitude. A modulation input provides for external jitter control, 
and an optional clock input provides for external data rate frequency control. 
.LP
	\(em
	\fIJitter receiver:\fR  | An instrument which demodulates and
measures the jitter present on a hierarchical clock or data signal. An 
output provides a voltage proportional to the demodulated jitter. 
.LP
	\(em
	\fILow\(hypass filter:\fR  | A circuit used to attenuate unwanted
spectral components above a given frequency.
.LP
	\(em
	\fIJitter measurement filter:\fR  | A circuit which attenuates
jitter spectral components outside a specified or desired
passband.
.LP
	\(em
	\fINetwork under test:\fR  | A circuit, system, or network that is
being tested using live traffic.
.LP
	\(em
	\fINoise source:\fR  | An instrument which generates a signal
having a near Gaussian amplitude distribution with a flat power
spectrum to approximately three times the half\(hypower bandwidth of
the retiming circuit.
.LP
	\(em
	\fISine wave generator:\fR  | A waveform generator which provides
a low distortion frequency and amplitude controlled sine
wave.
.LP
	\(em
	\fISpectrum analyzer:\fR  | An instrument which measures and
displays signal power as a function of frequency over a selected
frequency range. A tracking oscillator output provides an
adjustable amplitude swept frequency sinusoid which tracks the
instantaneous measurement frequency of the spectrum
analyzer.
.LP
	\(em
	\fIVoltmeter:\fR  | An instrument which measures DC, true rms, or
true peak\(hyto\(hypeak voltage as required. Here true peak\(hyto\(hypeak
voltage is defined as the difference between the most positive
and the most negative instantaneous voltages recorded during the
entire measurement interval.
.bp
.sp 2P
.LP
\fB4\fR 	\fBJitter tolerance measurement\fR 
.sp 1P
.RT
.PP
Jitter tolerance (also known as jitter accommodation) is defined in terms 
of the sinusoidal jitter amplitude which, when applied to an equipment 
input, causes a designated degradation of error performance. Jitter tolerance 
is a function of the amplitude and frequency of the applied jitter. 
.PP
Jitter tolerance requirements are specified in terms of jitter
templates which cover a specified sinusoidal amplitude/frequency region. 
Jitter templates represent the minimum amount of jitter an equipment \fImust 
accept\fR 
without causing the designated degradation of error performance (see Note).
.PP
The intended relationship of an equipment's actual tolerance to input jitter 
and its associated jitter tolerance template is illustrated in 
Figure\ 1.
.PP
\fINote\fR \ \(em\ In CCITT terminology, the jitter tolerance template
represents the \*Qlower limit of maximum tolerable input jitter\*U.
.RT
.sp 1P
.LP
4.1
	\fIActual tolerance\fR 
.sp 9p
.RT
.PP
The sinusoidal jitter amplitudes that an equipment actually
tolerates at a given frequency are defined as all amplitudes up to, but not
including, that which causes the designated degradation of error
performance.
.PP
The designated degradation of error performance may be expressed in
terms of either bit error ratio (BER) penalty or onset of errors criteria. 
The existence of two criteria arises because the input jitter tolerance 
of an 
individual digital equipment is primarily determined by the following two
factors:
.RT
.LP
	\(em
	The ability of the input clock recovery circuit to accurately
recover clock from a jittered data signal, possibly in the
presence of other degradations (pulse distortion, cross\(hytalk,
noise, etc.).
.LP
	\(em
	The ability of other components to accommodate dynamically
varying input data rates (e.g., pulse justification capacity and
synchronizer or desynchronizer buffer size in an asynchronous
digital multiplex).
.PP
The BER penalty criterion allows environment independent
determination of the decision circuit alignment jitter allocation, which is
critical for evaluating the first factor. A detailed discussion of the BER
penalty criterion may be found in References\ [6],\ [7]. The onset of errors
criterion is recommended for evaluating the second factor.
.sp 1P
.LP
4.1.1
	\fIBit error ratio penalty technique\fR 
.sp 9p
.RT
.PP
The bit error ratio (BER) penalty criterion for jitter tolerance
measurements is defined as the amplitude of jitter, at a given jitter
frequency, that duplicates the BER degradation caused by a specified signal 
to noise ratio (SNR) reduction. 
.PP
This technique is separated into two parts. Part one determines two
BER versus SNR reference points for the equipment under test. With zero 
jitter applied, noise is added to the signal, or the signal is attenuated, 
until a 
convenient initial BER is obtained. Then the noise, or signal attenuation, 
is decreased until the SNR at the decision circuit is increased the specified 
amount of dB (and consequently, the decision circuit is performing with an
improved BER). Part two uses the BER versus SNR reference points; at a given
frequency, jitter is added to the test signal until the BER returns to its
initially selected value. Since a known decision circuit eye width margin 
was established by the two BER versus SNR points, the added equivalent 
jitter is a true and repeatable measure of the decision circuit jitter 
tolerance 
performance. Part two of the technique is repeated for a sufficient number 
of frequencies such that the measurement accurately represents the continuous 
sinusoidal input jitter tolerance of the EUT over the applicable frequency
range. The test equipment must be able to produce a controlled jittered 
signal, a controlled SNR on the data stream, and measure the resulting 
BER from 
the\ EUT.
.PP
Figure\ 2 illustrates the test configuration for the BER penalty
technique. The equipment outlined in dashed lines is optional. The optional
frequency synthesizer is used to provide a more accurate determination of
frequencies utilized in the measurement procedure. This may be particularly
important for repeatability of measurements for some types of equipment; 
i.e., asynchronous digital multiplexes. The optional jitter receiver is 
used to 
verify the amplitude of generated jitter.
.bp
.RT
.LP
.rs
.sp 26P
.ad r
\fBFigure 2, (N), p.\fR 
.sp 1P
.RT
.ad b
.RT
.sp 1P
.LP
	\fIProcedure\fR 
.sp 9p
.RT
.LP
	i)
	Connect the equipment as shown in Figure\ 2. Verify proper
continuity and error\(hyfree operation.
.LP
	ii) 
	With no applied jitter, increase the noise (or attenuate
the signal) until at least 100 bit errors per second are
observed.
.LP
	iii)
	Record the corresponding BER and its associated SNR.
.LP
	iv) 
	Increase the SNR by the specified amount.
.LP
	v)
	Set the input jitter frequency as desired.
.LP
	vi) 
	Adjust the jitter amplitude until the BER returns to the
value recorded in step (iii).
.LP
	vii)
	Record the amplitude and frequency of the applied input
jitter, and repeat steps v)\ to\ vii) for a sufficient number of
frequencies to characterize the jitter tolerance
curve.
.sp 1P
.LP
4.1.2
	\fIOnset of errors technique\fR 
.sp 9p
.RT
.PP
The onset of errors criterion for jitter tolerance measurements is defined 
as the largest amplitude of jitter at a specified frequency that causes 
a cumulative total of more than\ 2 errored seconds, where these errored 
seconds have been summed over successive 30 seconds measurement intervals 
of increasing jitter amplitude. 
.PP
This technique involves setting a jitter frequency and determining the 
jitter amplitude of the test signal which causes the onset of errors criterion 
to be satisfied. Specifically, this technique requires: 
.RT
.LP
	1)
	isolation of the jitter amplitude \*Qtransition region\*U (in
which error\(hyfree operation ceases),
.LP
	2)
	 one errored second measurement, 30 seconds in duration, for each incrementally 
increased jitter amplitude from the beginning of this 
region, and
.LP
	3)
	 determination of the largest jitter amplitude for which the cumulative 
errored second count is no more than\ 2 errored seconds. 
.bp
.PP
The process is repeated for a sufficient number of frequencies
such that the measurement accurately represents the continuous sinusoidal 
input jitter tolerance of the EUT over the applicable jitter frequency 
range. The 
test equipment must be able to produce a controlled jittered signal and 
measure the resulting errored seconds caused by the jitter on the incoming 
signal. 
.PP
Figure\ 3 illustrates the test configuration for the onset of errors
technique. The optional frequency synthesizer is used to provide a more
accurate determination of frequencies utilized in the measurement procedure.
The optional jitter receiver is used to verify the amplitude of generated
jitter.
.RT
.LP
.rs
.sp 21P
.ad r
\fBFigure 3, (N), p.\fR 
.sp 1P
.RT
.ad b
.RT
.sp 1P
.LP
	\fIProcedure\fR 
.sp 9p
.RT
.LP
	i)
	Connect the equipment as shown in Figure\ 3. Verify proper
continuity and error\(hyfree operation.
.LP
	ii)
	 Set the input jitter frequency as desired, and initialize the jitter 
amplitude to 0\ UI peak\(hypeak. 
.LP
	iii)
	Increase the jitter amplitude in gross increments to
determine the amplitude region where error\(hyfree operation ceases. Reduce the
jitter amplitude to its level at the beginning of this region.
.LP
	iv)
	 Record the number of errored seconds that occur over a 30 second measurement 
interval. Note that the initial measurement must be\ 0 
errored seconds.
.LP
	v)
	 Increase the jitter amplitude in fine increments, repeating step\ iv) 
for each increment, until the onset of errors criterion is satisfied. 
.LP
	vi)
	 Record the indicated amplitude and frequency of the applied input jitter, 
and repeat steps\ ii) to\ iv) for a sufficient number of 
frequencies to characterize the jitter tolerance curve.
.sp 1P
.LP
4.2
	\fIJitter tolerance template compliance\fR 
.sp 9p
.RT
.PP
Equipment jitter tolerance is specified with jitter tolerance
templates. Each template defines the region over which the equipment must
operate without suffering the designated degradation of error performance. 
The difference between the template and actual equipment tolerance curve 
represents the operating jitter margin, illustrated in Figure\ 1. 
.bp
.PP
The template compliance measurement is performed by setting the jitter 
frequency and amplitude to the template value, and observing that the 
designated degradation of error performance does not occur.
.PP
A sufficient number of template points are measured to assure
compliance over the entire frequency range of the template.
.PP
Figure 2 or 3, as applicable, illustrates the test configuration for the 
jitter tolerance template compliance technique. 
.RT
.sp 1P
.LP
	\fIProcedure\fR 
.sp 9p
.RT
.LP
	i)
	 Connect the equipment as described in \(sc\ 4.1.1 or 4.1.2, as applicable. 
Verify proper continuity and error\(hyfree operation. 
.LP
	ii)
	Set the jitter amplitude and frequency to a template
point.
.LP
	iii)
	When the onset of errors technique is used, confirm
that\ 0 errored seconds occur. When the BER penalty technique is used, 
confirm that the designated degradation of error performance is not reached. 
.LP
	iv)
	Repeat steps ii) and iii) for a sufficient number of
template points to verify jitter tolerance template compliance.
.sp 2P
.LP
\fB5\fR 	\fBJitter transfer characteristic measurement\fR 
.sp 1P
.RT
.PP
The jitter transfer characteristic of an individual digital
equipment is defined as the ratio of the output jitter to the applied input
jitter as a function of frequency.
.PP
If the relationship between the jitter appearing at the input and
output ports of a digital equipment can be described in terms of a linear
process (a process which is both additive and homogeneous), the term \*Qjitter
transfer function\*U is used. The relationship between jitter appearing at the
input and output ports of some types of digital equipment cannot be described 
in terms of a jitter transfer function. In such cases, different measurement 
techniques may be necessary to obtain meaningful results.
.RT
.sp 1P
.LP
5.1
	\fILinear processes\fR 
.sp 9p
.RT
.PP
Jitter transfer measurements are commonly required for clock
recovery circuits and desynchronizer phase smoothing circuits. Measurement 
of the jitter transfer function of a linear clock recovery circuit is generally 
straightforward. However, measurement of the jitter transfer function of a
linear desynchronizer phase smoothing circuit requires specialized techniques 
because it is embedded in a non\(hylinear asynchronous digital multiplex. 
.RT
.sp 1P
.LP
5.1.1
	\fIClock recovery circuit\fR 
.sp 9p
.RT
.PP
Clock recovery circuits are an essential component of individual
digital equipment input ports. Of particular interest is the jitter transfer
function of clock recovery circuits which dominate the transfer of jitter 
from the input to output ports. Characterization of linear clock recovery 
circuits embedded in non\(hylinear equipment (i.e., asynchronous digital 
multiplexes) are not addressed because they typically do not dominate the 
overall equipment 
jitter transfer characteristic.
.RT
.sp 1P
.LP
5.1.1.1
	\fIBasic technique\fR 
.sp 9p
.RT
.PP
This technique involves applying swept sinusoidal jitter at a
fixed tolerable amplitude over a selected frequency range to the EUT, and
observing the output jitter amplitude over the applied frequency range. The
process is repeated for a sufficient number of frequency ranges to characterize 
the jitter transfer function of the EUT. 
.PP
Specifically, this technique utilizes a spectrum analyzer to set a
jitter frequency range and corresponding tolerable jitter amplitude. Initially, 
the EUT is bypassed to establish a\ 0\ dB amplitude reference trace for 
the test equipment. The EUT is then re\(hyconnected, and the\ 0\ dB amplitude 
reference trace subtracted from the overall jitter transfer measurement 
to obtain the EUT 
jitter transfer function. Use of a spectrum analyzer with a tracking oscillator 
output is required to determine the input jitter frequency and amplitude 
while making a narrow\(hyband measurement of the output jitter. To achieve 
a high degree of accuracy, the spectrum analyzer bandwidth must be sufficiently 
narrow to 
obtain the desired amplitude resolution and dynamic range in each frequency
band measured. For example, to verify less than 0.1\ dB peaking, and a 
20\ dB per decade roll\(hyoff from 350\ Hz to 20\ kHz, a spectrum analyzer 
with 0.1\ dB 
resolution, 3\ Hz bandwidth, and 40\ dB dynamic range may be required.
.bp
.PP
Figure\ 4 illustrates the test configuration for the jitter transfer
function measurement. The optional frequency synthesizer may be used to 
provide a more accurate determination of frequencies utilized in the measurement 
procedure.
.RT
.LP
.rs
.sp 21P
.ad r
\fBFigure 4, (N), p.\fR 
.sp 1P
.RT
.ad b
.RT
.sp 1P
.LP
	\fIProcedure\fR 
.sp 9p
.RT
.LP
	i)
	 Perform a jitter tolerance measurement of the EUT over the desired frequency 
range, as described in\ \(sc\ 4. 
.LP
	ii)
	 Connect the equipment as shown in Figure\ 4, bypassing the EUT. Verify 
proper continuity, linearity, and error\(hyfree operation. 
.LP
	iii)
	Set the frequency range on the spectrum analyzer as
desired. Adjust the tracking oscillator output level on the spectrum analyzer 
to produce a tolerable jitter amplitude over the selected frequency range, 
which is large enough to ensure adequate measurement accuracy, yet sufficiently 
small to preserve linear operation. 
.LP
	iv)
	Setting the spectrum analyzer bandwidth as narrow as
feasible, sweep the desired frequency range, and record the 0\ dB amplitude
reference trace of the test equipment. (Setting a narrow spectrum analyzer
bandwidth may allow a reduction in applied jitter amplitude with no loss in
measurement accuracy.)
.LP
	v)
	Reconnect the EUT as shown in Figure\ 4. Verify proper
continuity, linearity, and error\(hyfree operation.
.LP
	vi)
	 Use the spectrum analyzer to sweep the selected frequency range and record 
the magnitude of the overall (test equipment and EUT) jitter transfer function. 
.LP
	vii)
	 To obtain the EUT jitter transfer function, substract the 0\ dB amplitude 
reference trace from the overall jitter transfer function 
recorded in step\ vi).
.LP
	viii)
	Repeat steps\ i) to\ vii) for a sufficient number of
frequency ranges to characterize the overall frequency range of
interest.
.bp
.sp 1P
.LP
5.1.2
	\fIDesynchronizer phase smoothing circuit\fR 
.sp 9p
.RT
.PP
In general, a non\(hylinear process characterizes the relationship
between the jitter appearing at the input and output ports of an asynchronous 
digital multiplex. However, most phase smoothing circuits are intended 
to 
operate linearly, and therefore may have a transfer function associated with
them. Two techniques have been developed which enable the determination 
of the jitter transfer function for a linear desynchronizer phase\(hy 
smoothing
circuit
using standard multiplex interfaces. The first technique utilizes interfaces 
at the multiplexer low\(hyspeed input and the demultiplexer low\(hyspeed 
output. The 
second technique utilizes the interfaces at the demultiplexer high\(hyspeed 
input and low\(hyspeed output. The second technique utilizes the interfaces 
at the 
demultiplexer high\(hyspeed input and low\(hyspeed output.
.RT
.sp 1P
.LP
5.1.2.1
	\fIMultiplex technique\fR 
.sp 9p
.RT
.PP
This technique attempts to \*Qlinearize\*U the multiplexing process by 
applying appropriate constraints to the applied input jitter amplitude 
and 
frequency. Sinusoidal jitter of a selected amplitude and frequency is applied 
to the multiplexer low\(hyspeed output is observed at the applied frequency. 
The process is repeated for a sufficient number of frequencies to characterize 
the desynchronizer jitter transfer function. Specifically, when sinusoidal 
jitter modulates the phase of the input signal to one of the multiplexer 
low\(hyspeed 
inputs, the jitter spectrum appearing at the corresponding tributary outputs, 
in addition to containing other waiting time jitter components at discrete 
locations throughout the spectrum, contains a discrete component at the
frequency of the input jitter. This technique involves making the amplitude 
of the input jitter sufficiently large to ensure that this discrete component 
in the output jitter spectrum at the applied frequency dominates the other 
.PP
waiting time jitter components in the measurement bandwidth. However, it 
should not be so large as to saturate the multiplexer stuffing mechanism 
(onset of 
saturation). The smallest magnitude of frequency deviation,
\fIf\fR (\fIt\fR ), which causes
onset of saturation is determined from the smaller magnitude of:
\v'6p'
.RT
.sp 1P
.ce 1000
\fIf\fR (\fIt\fR ) = \fIf\fR \fI\fI\d\fIs\fR\\d\fIc\fR\u\(em \fIf\fR \fI\fI\d\fIn\fR\\d\fIo\fR\\d\fIm\fR\u 
.ce 0
.sp 1P
.ce 1000
\fIf\fR (\fIt\fR ) = \(em\fIf\fR \fI\fI\d\fIm\fR\u+ \fIf\fR \fI\fI\d\fIs\fR\\d\fIc\fR\u\(em 
\fIf\fR \fI\fI\d\fIn\fR\\d\fIo\fR\\d\fIm\fR\u
.ce 0
.sp 1P
.LP
.sp 1
.LP
where
.LP
	 \fIf\fR \fI\fI\d\fIs\fR\\d\fIc\fR\u represents the multiplexer average 
synchronous data 
bit read clock rate,
.LP
	\fIf\fR \fI\fI\d\fIm\fR\u	represents the maximum rate at which pulses can be
stuffed into an incoming pulse stream, and
.LP
	 \fIf\fR \fI\fI\d\fIn\fR\\d\fIo\fR\\d\fIm\fR\u refers to the nominal incoming 
line rate. 
.PP
To achieve a high degree of accuracy, the spectrum analyzer
bandwidth must be sufficiently narrow to obtain the desired amplitude
resolution and dynamic range in each frequency band measured (see \(sc\ 
5.1.1.1). It is also assumed that the transfer function of the multiplexer 
low\(hyspeed 
input clock recovery circuit does not alter the applied jitter in the frequency 
range of interest. 
.PP
Figure 4 illustrates the test configuration for the jitter transfer
function measurement. The optional frequency synthesizer may be used to 
provide a more accurate determination of frequencies utilized in the measurement 
procedure.
.RT
.sp 1P
.LP
	\fIProcedure\fR 
.sp 9p
.RT
.LP
	i)
	Perform a jitter tolerance measurement over the desired
frequency range.
.LP
	ii)
	 Connect the equipment as shown in Figure\ 4, bypassing the EUT. Verify 
proper continuity, linearity, and error\(hyfree operation. 
.LP
	iii)
	Manually set the test frequency on the spectrum analyzer.
.LP
	iv)
	 Adjust the tracking oscillator output level on the spectrum analyzer 
to produce the largest tolerable jitter amplitude which will not cause 
onset of saturation (as defined in this paragraph) at the selected 
frequency.
.LP
	v)
	 Set the spectrum analyzer bandwidth as narrow as feasible, and record 
the 0\ dB amplitude transfer reference level of the test 
equipment.
.LP
	vi)
	Reconnect the EUT as shown in Figure\ 4. Verify proper
continuity and error\(hyfree operation.
.bp
.LP
	vii) 
	 Record the magnitude of the overall (test equipment and EUT) jitter transfer 
function. Averaging is generally required to remove the 
effects of waiting time jitter on the measurement.
.LP
	viii)
	To obtain the magnitude of the EUT jitter transfer
function, substract the 0\ dB amplitude transfer reference level from the
overall magnitude obtained in step\ vii).
.LP
	ix)
	Repeat steps iii) \(em viii) for a sufficient number of
frequencies to characterize the jitter transfer function of
the EUT.
.sp 1P
.LP
5.1.2.2
	\fIDemultiplexer technique\fR 
.sp 9p
.RT
.PP
This technique involves applying sinusoidal jitter of a selected
amplitude and frequency to the demultiplexer high\(hyspeed input, and observing
the jitter amplitude at the demultiplexer low\(hyspeed output at the applied
frequency. The process is repeated for a sufficient number of frequencies to
characterize the desynchronizer jitter transfer function. Specifically, when
sinusoidal jitter modulates the phase of the input signal to the demultiplexer, 
the output jitter spectrum contains a discrete component at the frequency 
of 
the input jitter, in addition to the intrinsic waiting time jitter components 
already present. This technique involves making the amplitude of the applied 
input jitter sufficiently large to ensure that its contribution to the 
output jitter spectrum at the applied frequency dominates that of the waiting 
time 
jitter, but does not exceed the demultiplexer input jitter tolerance. It is
also assumed that the transfer function of the demultiplexer high\(hyspeed 
input clock recovery circuit does not alter the applied jitter in the frequency 
range of interest. 
.PP
Figure 4 illustrates the test configuration for the jitter transfer
function measurement. It should be emphasized that the following procedure
\fIcannot calibrate out the effects of the low\(hyspeed receive circuitry 
contained\fR \fIin the jitter receiver functional block component\fR , 
and therefore requires 
that this circuitry has flat response.
.PP
It should be noted that the digital signal applied to the high\(hyspeed 
input of the demultiplexer must contain framing information to allow proper 
operation of the equipment under test. \*QFramed\*U signals can either be taken
from an appropriate digital signal generator or may come from the corresponding 
digital multiplexer. In the latter case, a transparent jitter modulator 
has to be inserted between the high\(hyspeed multiplexer output and the 
demultiplexer 
input. The jitter modulator superimposes jitter on the jitter\(hyfree signal
coming from the multiplexer.
.RT
.sp 1P
.LP
	\fIProcedure\fR 
.sp 9p
.RT
.LP
	i)
	 Follow the procedure provided in \(sc\ 5.1.1.1 using Figure\ 4, scaling 
the applied jitter in unit intervals (UI) by the ratio of the 
demultiplexer high\(hyspeed input to low\(hyspeed output data rates.
.sp 1P
.LP
5.2
	\fINon\(hylinear process\fR 
.sp 9p
.RT
.PP
This area requires further study.
.RT
.sp 2P
.LP
\fB6\fR 	\fBOuput jitter measurement\fR 
.sp 1P
.RT
.PP
Output jitter measurements fall within two categories:
.RT
.LP
	1)
	network output jitter at hierarchical interfaces, and
.LP
	2)
	intrinsic jitter generated by individual digital
equipment.
.PP
Measurements of output jitter may be in terms of rms and
peak\(hyto\(hypeak amplitudes over designated frequency ranges, and may require
statistical characterization.
.PP
Output jitter measurements utilize either live traffic or controlled data 
patterns. 
.RT
.sp 1P
.LP
6.1
	\fILive traffic\fR 
.sp 9p
.RT
.PP
Output jitter measurements at network hierarchical interfaces
typically use a live traffic signal. For pre\(hyservice testing, in which
controlled data patterns are used, see \(sc\ 6.2. This technique involves
demodulating the jitter from the live traffic at the output of a network
interface, selectively filtering the jitter, and measuring the true rms 
or true peak\(hyto\(hypeak amplitude of the jitter over the specified measurement 
time 
interval.
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.PP
Figure\ 5 illustrates the test configuration for the live traffic
technique. The optional spectrum analyzer allows observation of the output
jitter frequency spectrum.
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\fBFigure 5, (N), p.\fR 
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	\fIProcedure\fR 
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	i)
	Connect the equipment as shown in Figure\ 5. Verify proper
continuity and error\(hyfree operation.
.LP
	ii) 
	Select the desired jitter measurement filter and measure
the filtered output jitter, recording the true peak\(hyto\(hypeak
jitter amplitude that occurs during the specified measurement
time interval.
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	iii)
	Repeat step ii) for all desired jitter measurement
filters.
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6.2
	\fIControlled data patterns\fR 
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.PP
Measurement of intrinsic jitter in individual digital equipment
requires the application of controlled data patterns. Controlled data patterns 
are generally applicable in laboratory, factory, and out\(hyof\(hyservice 
situations. The \*Qbasic technique\*U, described below, details how such 
measurements may be 
performed.
.PP
Where it is desirable to obtain more detailed information regarding
output jitter power (specifically, jitter generated in digital regenerators), 
jitter may be further categorized in terms of random and systematic components. 
The primary reasons for distinguishing between random and systematic jitter 
are to enable the comparison of measurement results with theoretical computations, 
and to refine regenerator design. The \*Qenhanced technique\*U [6] describes 
how 
random and systematic jitter may be measured.
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6.2.1
	\fIBasic technique\fR 
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This technique is indentical to that described in\ \(sc\ 6.1, except for 
the application of an unjittered controlled data pattern to the EUT. In 
Figure\ 5, the optional frequency synthesizer may be used to provide a more
accurate determination of frequencies utilized in the measurement
procedure.
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	\fIProcedure\fR 
.sp 9p
.RT
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	i)
	Connect the equipment as shown in Figure\ 5, using the
digital signal generator to provide an unjittered controlled data pattern 
to the EUT. Verify proper continuity and error\(hyfree operation. 
.LP
	ii)
	Select the desired jitter measurement filter and measure
the filtered output jitter, recording the true peak\(hyto\(hypeak jitter 
amplitude 
that occurs during the specified measurement time interval.
.LP
	iii)
	Repeat step ii) for all desired jitter measurement
filters.
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	\fBReferences\fR 
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.RT
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[1]
	CCITT Recommendation \fIVocabulary of digital transmission and\fR 
\fImultiplexing, and pulse code modulation (PCM) terms\fR , Volume\ III,
Rec.\ G.701.
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[2]
	CCITT Recommendation \fIDigital line sections at 1544 kbit/s\fR ,
\fITest sequencies for jitter measurements on digital line sections\fR ,
Red\ Book, Volume\ III, Rec.\ G.911, Annex\ A. ITU, Geneva, 1984.
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[3]
	Hucket (P.): Performance evalution in an ISDN \(em Digital transmission
impairments, \fIRadio and Electronic Engineer\fR , Volume\ 54, No.\ 2, 
pp.\ 97\(hy106, 
February\ 1984.
.LP
[4]
	CCITT Recommendation \fIPhysical/electrical characteristics of\fR 
\fIhierarchical digital interfaces\fR , Volume\ III, Rec.\ G.703.
.LP
[5]
	T1X1.4/85\(hy031 \fIProposed draft American national standard for DS1,\fR 
\fIDS1C and DS3 levels of the digital hierarchy\fR .
.LP
[6]
	Trischitta (P.R.): Jitter accumulation in fiber optic systems,
\fIRutgers\fR , The State University of New Jersey, May,\ 1986.
.LP
[7]
	Trischitta (P.R.), Sannuti (P.): The jitter tolerance of fiber optic
regenerators, \fIIEEE Transactions on Communications\fR , Vol.\ 35, No.\ 
12, pp. 
1303\(hy1308, December,\ 1987.
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